Switching power supply circuit

ABSTRACT

A switching power supply circuit is disclosed which allows selective use of a switching element having a comparatively low voltage withstanding property and a low current capacity and can be produced in a low lost and with a small size. Upon starting of power supply, the switching frequency of the switching element is raised by a soft starting circuit to increase a period of time until a secondary side DC output voltage reaches a steady level thereof As a result, the levels of a resonance voltage and a collector current obtained when the switching element starts its switching operation are suppressed.

TECHNICAL FIELD

This invention relates to a switching power supply circuit which can beincorporated as a power supply in various electronic apparatus.

BACKGROUND ART

A switching power supply circuit which adopts a switching converter inthe form of, for example, a flyback converter or a forward converter iswidely known. Because switching converters of these types employ arectangular waveform signal to control a switching operation, theseswitching converters are limited in the amount of switching noise theycan suppress. It is also known that these switching converters also arelimited in power conversion efficiency because of their operationcharacteristics.

Thus, various switching power supply circuits that employ resonance typeconverters have been proposed by the assignee of the presentapplication. A resonance type converter is advantageous in that a highpower conversion efficiency can be readily obtained while maintaininglow noise characteristics because the waveform controlling the switchingoperation is a sine waveform. This resonance type converter is alsoadvantageous in that it can be simply formed from a comparatively smallnumber of parts.

FIG. 7 shows an example of a switching power supply circuit which hasbeen previously proposed by the assignee of the present application.Referring to FIG. 7, a power supply circuit 700 includes a voltageresonance type converter including a single switching element Q1 thatperforms a switching operation in a self-excited manner in accordancewith a single end system.

Switching power supply circuit 700 includes a rectifier smoothingcircuit for receiving a commercial ac power supply (ac input voltageVAC) and producing a DC input voltage. The rectifier smoothing circuitis formed as a full-wave rectifier circuit, comprising a bridgerectifier circuit Di and a smoothing capacitor Ci. The rectifiersmoothing circuit produces a rectified smoothed voltage Ei of a levelequal to the ac input voltage VAC. Further, an inrush current limitationresistor Ri is interposed in a rectifier current path of the rectifiersmoothing circuit in order to suppress any initial inrush current spikefrom flowing into a smoothing capacitor Ci, for example, when an initialpower supply is provided to the circuit. Further, in power supplycircuit 700 an AC switch SW is interposed in the commercial ac powersupply line. AC switch SW is switched on/off to start/stop flow of powerto power supply circuit 700.

A voltage resonance type switching converter is provided in power supplycircuit 700. This switching converter, as described above, has aself-exciting construction including switching element Q1. In thisinstance, switching element Q1 may be formed of a bipolar transistor(BJT: junction transistor) having a high voltage withstanding property.As is shown in FIG. 7, the base of switching element Q1 is connected tothe positive electrode side of smoothing capacitor Ci (rectifiedsmoothed voltage Ei) through a starting resistor RS so that the basecurrent upon start-up of the circuit may be obtained from the rectifiersmoothing circuit. Further, a resonance circuit adapted to be driven ina self-excited oscillation state is connected between the base ofswitching element Q1 and a primary side ground 710. The resonancecircuit is formed from the series circuit connection of an inductioncharacteristic LB of a detection driving winding NB, a resonancecapacitor CB, and a base current limiting resistor RB.

A damper diode DD is interposed between the base of the switchingelement Q1 and the negative electrode (set at a primary side ground) ofsmoothing capacitor Ci and forms a path for damper current which flowswhen the switching element Q1 is switched off. The collector ofswitching element Q1 is connected to an end of a primary winding N1 ofan insulating converter transformer PIT. The emitter of switchingelement Q1 is grounded.

A parallel resonance capacitor Cr is connected in parallel between thecollector and the emitter of switching element Q1. Parallel resonancecapacitor Cr forms, based on a capacitance thereof and a leakageinductance L1 of primary winding N1 of insulating converter transformerPIT, a primary side parallel resonance voltage resonance type convertercircuit. Although detailed description is omitted here, when theswitching element Q1 is off, a voltage resonance type operation isobtained by an action of the parallel resonance circuit which causes thevoltage Vcp across resonance capacitor Cr to actually exhibit a pulsewave of a sine waveform.

An orthogonal control transformer PRT shown in FIG. 7 is a saturatablereactor on which a detection winding ND, a drive winding NB and acontrol winding NC are wound. Orthogonal control transformer PRT isprovided for driving switching element Q1 and controlling an outputvoltage to be constant. Though not shown, orthogonal control transformerPRT is formed with a structure wherein a three dimensional core isformed such that two double channel-shaped cores each having fourmagnetic legs are joined to each other at the ends of the magnetic legsthereof. Detection winding ND and Drive winding NB are wound in the samewinding direction around two predetermined ones of the magnetic legs ofthe three dimensional core, and the control winding NC is wound aroundtwo predetermined ones of the magnetic legs of the three dimensionalcore such that the winding direction thereof is orthogonal to detectionwinding ND and the drive winding NB.

Detection winding ND of orthogonal control transformer PRT (frequencyvariation means) is interposed in series between the positive electrodeof smoothing capacitor Ci and primary winding N1 of insulating convertertransformer PIT so that a switching output of the switching element Q1is transmitted to detection winding ND through primary winding N1 . Inorthogonal control transformer PRT, the switching output obtained indetection winding ND is excited in driving winding NB via transformercoupling, and consequently, an alternating drive voltage is generated indriving winding NB. The drive voltage is output as drive current fromthe series resonance circuit (NB and CB), which forms the self-excitedoscillation drive circuit, to the base of switching element Q1 throughbase current limiting resistor RB. Consequently, switching element Q1performs a switching operation at a switching frequency determined bythe resonance frequency of the series resonance circuit (NB and CB).Insulating converter transformer PIT transmits a switching output of theswitching element Q1 to the secondary side thereof, including secondarywinding N₂.

Referring next to FIG. 8, the structure of insulating convertertransformer PIT will be described. Insulating converter transformer PITincludes an EE-shaped core which includes a pair of E-shaped cores CR1and CR2 made of, for example, a ferrite material and coupled with eachother such that magnetic legs thereof are opposed to each other. Aprimary winding N1 and a secondary winding N2 are wound separately fromeach other on the central magnetic legs of the EE-shaped core using asplit bobbin B. As seen from FIG. 8, a gap G is formed between thecentral magnetic legs of the EE-shaped core. Consequently, a loosecoupling having a required coupling coefficient can be obtained. The gapG can be formed by providing the central magnetic legs of the E-shapedcores CR1 and CP2 shorter than the other two outer magnetic legs. Thecoupling coefficient k in this instance is, for example, k0 0.85 whichis a coupling coefficient of a loose coupling. Consequently, asaturation condition is less likely to be obtained as much.

Referring back to FIG. 7, one end of primary winding N1 of insulatingconverter transformer PIT is connected to the collector of the switchingelement Q1. The other end of primary winding N1 is connected to thepositive electrode of smoothing capacitor Ci (rectified smoothed voltageEi) through a series connection of detection winding ND.

On the secondary side of insulating converter transformer PIT, analternating voltage induced by primary winding Ni appears in secondarywinding N2. A secondary side parallel resonance capacitor C2 isconnected in parallel to secondary winding N2. A parallel resonancecircuit is therefore formed from a leakage inductance L2 of secondarywinding N2 and a capacitance of secondary side parallel resonancecapacitor C2. The alternating voltage induced in secondary winding N2 isconverted into a resonance voltage by the parallel resonance circuit. Inshort, a voltage resonance operation is obtained on the secondary side.

Thus, in the power supply circuit of FIG. 7, a parallel resonancecircuit for generating a voltage resonance type switching operation isprovided on the primary side of insulated converter transformer PIT, anda parallel resonance circuit for obtaining a full-wave rectificationoperation (voltage resonance operation) is provided on the secondaryside of insulated converter transformer PIT. It is to be noted that, inthe present specification, a switching converter of a construction whichincludes resonance circuits for both of the primary side and thesecondary side in this manner is suitably referred to as “compositeresonance type switching converter”.

In the parallel resonance circuit on the secondary side of insulatedconverter transformer PIT formed as described above, center taps areprovided for secondary winding N2, and rectifier diodes D01, D02, D03and D04 and smoothing capacitors C01 and C02 are connected in such amanner as shown in FIG. 7 to provide two full-wave rectifier circuits; afirst full-wave rectifier circuit including rectifier diodes D01 and D02and smoothing capacitor C01, and a second full-wave rectifier circuitincluding rectifier diodes D03 and D04 and smoothing capacitor C02.

The first full-wave rectifier circuit composed of rectifier diodes D01and D02 and smoothing capacitor C01 receives a resonance voltagesupplied from the secondary side parallel resonance circuit and producesa DC output voltage E01. The second full-wave rectifier circuit composedof rectifier diodes D03 and D04 and smoothing capacitor C02 similarlyreceives the resonance voltage supplied from the secondary side parallelresonance circuit and produces a DC output voltage E02.

The DC output voltage E01 and the DC output voltage E02 are also inputto a control circuit 1. Control circuit 1 utilizes the DC output voltageE01 as a detection voltage and utilizes the DC output voltage E02 as anoperation power supply therefor.

In insulating converter transformer PIT, the mutual inductance M betweenthe inductance L1 of primary winding N1 and the inductance L2 ofsecondary winding N2 may have a value +M or a value −M depending uponthe relationship between the polarities (winding directions) of primarywinding N1 and secondary winding N2 and the connection of rectifierdiodes D0 (D01 and D02). For example, if the components are connected ina configuration as shown in FIG. 9A, then the mutual inductance is +M.If the components are connected in a configuration as shown in FIG. 9B,then the mutual inductance is −M.

If this is examined in connection with operation of the secondary sideof the circuit of FIG. 7, for example, the operation that rectifiedcurrent flows through the rectifier diode D01 (D03) when the alternatingvoltage obtained at the secondary winding N2 has the positive polaritycan be regarded as an operation mode of +M (forward mode). On thecontrary, the operation that rectified current flows through therectifier diode D02 (D04) when the alternating voltage obtained at thesecondary winding N2 has the negative polarity can be regarded as anoperation mode of −M (flyback mode). In other words, the power supplycircuit of FIG. 7 operates in the +M/−M mode of the mutual inductanceeach time the alternating voltage obtained at the secondary windingbecomes positive/negative.

In the power supply circuit having the construction described above,power output from the secondary winding side parallel resonance circuitmay be increased based upon the supply of power thereto. In thisinstance, for example, if a full-wave rectifier circuit is connected tothe secondary winding side parallel resonance circuit as in the circuitshown in FIG. 7, rectified current flows alternately in both of the+M/−M operation modes of the mutual inductance as described above. Inother words, a rectified output from the circuit is obtained within bothof the periods within which the alternating voltage is positive andnegative. By this described operation, supplied power increases, andalso the rate of reaching the maximum load power increases.

The construction for obtaining the full-wave rectification operationillustrated in FIGS. 9A and 9B is realized by forming gap G ininsulating converter transformer PIT of FIG. 8 (and FIG. 7) to obtain aloose coupling of a predetermined coupling coefficient to establish acondition wherein the insulating converter transformer is less liable toreach a saturation condition. For example, where gap G is not providedin the insulating converter transformer PIT, there is a high degree ofpossibility that, upon performance of flyback operation, insulatingconverter transformer PIT may be put into a saturation condition andoperate abnormally. The full-wave rectification operation describedabove is therefore less likely to be properly preformed in such asaturation condition.

Control circuit 1, shown in FIG. 7, varies the control current (DCcurrent) level supplied to control winding NC in response to the levelof the DC output voltage (E01) of the secondary winding side. Thissupplied current level is adjusted to variably control the inductance LBof drive winding NB wound on the orthogonal control transformer PRT in afeedback control loop-type setup. Consequently, the resonance conditionof the series resonance circuit formed of inductance LB of the drivewinding NB varies. This variation in turn varies the switching frequencyof the switching element Q1 as hereinafter described with reference toFIG. 11, and the secondary winding side DC output voltage is stabilizedby the variation of the switching frequency of switching element Q1.

In the power supply circuit shown in FIG. 7, in order to vary theswitching frequency, the period within which switching element Q1 is offis fixed whereas the period within which switching element Q1 is on isvariably controlled. While employing a constant voltage control, thecontrol operation for the power supply circuit operates to variablycontrol the switching frequency of switching element Q1 to performresonance impedance control for the switching output, and simultaneouslyperforms continuity angle control (PWM control) of the switching elementin a switching period. This composite control operation is realized witha single control circuit system. Here, the switching frequency controlis performed such that, when the secondary side output voltage rises asa result of, for example, decreasing of the load thereon, the switchingfrequency is raised to suppress the secondary winding side output power.

During use, AC switch SW in power supply circuit 700 shown in FIG. 7 isswitched from an off-state to an on-state. Charging current then flowsinto smoothing capacitor Ci through inrush current limiting resistor Riand bridge rectifier circuit Di so that charging of smoothing capacitorCi is preformed. This charging continues until the rectified smoothedvoltage Ei which is a voltage across smoothing capacitor Ci rises to alevel corresponding to the ac input voltage level. Then, when startingcurrent is supplied from the rectified smoothed voltage Ei to the baseof switching element Q1 through starting resistor RS, switching elementQ1 is turned on to begin oscillation thereof. Thereafter, the switchingelement Q1 performs a switching operation according to the oscillationthereof.

When this process begins, and Q₁ is in the on state, the secondarywinding side of insulated converter transformer PIT has a low impedance.An excessively high charging current flows into smoothing capacitors C01and C02 in which no charge has yet been accumulated. In other words, thesecondary winding side DC output voltage is in a transient state until asteady level is obtained, and in this state, control of the switchingfrequency by the control circuit 1 is not properly preformed. At thistime, the switching element Q1 performs a switching operation with thelowest switching frequency which depends upon the time constant of theself-excited oscillation drive circuit (CB and NB).

Therefore, upon startup, the period of switching element Q1 becomes longin accordance with operation of the PWM control described above. Theresonance voltage pulse Vcp generated by switching element Q1 within aperiod within which the switching element Q1 is off therefore becomesexcessively high. Consequently, because of excessive inrush currentflowing to smoothing capacitors C01 and C02 on the secondary windingside of insulated converter transformer PIT, excessive collector currentIcp flows to switching element Q1 in the primary side from the smoothingcapacitor Ci through detection winding ND and primary winding N1.

Because a resonance voltage pulse Vcp and collector current Icp ofexcessively high levels are generated in such a manner upon startup ofthe circuit, switching element Q1 must be selected to have a highvoltage withstanding property and a high current flow capacity towithstand them. A switching element having such characteristics isexpensive and large in size, and therefore becomes an obstacle toreducing the cost and miniaturizing the size of the circuit.

Therefore, it is also a common practice to provide an over currentlimiting circuit in order to allow use of a switching element having acomparatively low voltage withstanding property and current capacity. Aconstruction of a power supply circuit of this type is shown in FIG. 10.It is to be noted that in FIG. 10, like reference characters to those ofFIG. 7 denote like elements, and description thereof is omitted hereinto avoid redundancy.

The circuit shown in FIG. 10 includes an over current limiting circuit10 in addition to the circuit described hereinabove with reference toFIG. 7. Over current limiting circuit 10 includes a current detectionresistor RE and a series connection circuit of voltage dividingresistors R11 and R12 interposed in parallel to each other between theemitter of switching element Q1 and the primary winding side ground. Thebase of a transistor Q2, used for conduction control, is connected to ajunction between voltage dividing resistors R11 and R12.

The collector of transistor Q2 is connected to the base of switchingelement Q1 through a diode D10. The anode of the diode D10 is connectedto the base of switching element Q1. The cathode of diode D10 isconnected to the collector of transistor Q2. The emitter of transistorQ2 is grounded to the primary winding side ground. A capacitor C10, forexample, for noise absorption, is connected in parallel between thecollector and the emitter of transistor Q2.

In the power supply circuit having the construction shown in FIG. 10,collector current Icp which is generated upon start-up flows from thecollector through the emitter of switching element Q1 and is detected bycurrent detection resistor RE and the series connection of voltagedividing resistors R11 and R12. A current level corresponding to avoltage dividing ratio of voltage dividing resistors R11 and R12 flowsto the base of transistor Q2. When the level of the collector currentIcp of transistor switching element Q₁ becomes higher than a certainpredetermined level and the base current amount of transistor Q2 alsobecomes higher than a certain predetermined level, transistor Q2 isrendered conductive thereby to render diode D10 conductive.

When diode D10 and transistor Q2 are conductive, forward current flowingto the base of transistor Q2 flows to the primary winding side groundthrough diode D10 and transistor Q2 (collector-emitter). In short, thecollector current Icp is limited by limiting excessive base current totransistor Q2 upon start-up.

While excessive current can be limited in such a manner, the powersupply circuit of FIG. 10 still has the following problems. First,because current detection resistor RE is connected in series to theemitter of switching element Q1, when a load placed thereon is heavy,high power loss occurs and the power conversion efficiency deteriorates.Further, when the power supply circuit shown in FIG. 10 is to beactually designed, a margin must be provided against a malfunction whenthe ac input voltage VAC rises high or against a malfunction whicharises from a variation of some component of the over current limitingcircuit, even upon steady operation, so that the malfunction does notoccur.

FIGS. 11A to 11D are waveform diagrams illustrating operation of thepower supply circuit of FIG. 10 which is designed to include a marginagainst a malfunction. Particularly, FIGS. 11A and 11B show waveforms ofthe resonance voltage Vcp and the collector current Icp, respectively,when the maximum load power Pomax is 150 W and the ac input voltage VACis 120 V. FIGS. 11C and 11D show waveforms of the resonance voltage Vcpand the collector current Icp, respectively, in a steady statecondition.

The switching frequency of switching element Q1 is controlled so as tobe lower in a maximum load power operation condition than in a steadystate operation. Switching element Q1 is thus controlled such that,while the period TOFF within which the switching element Q1 is off isfixed, the period TON within which the switching element Q1 is onincreases. Actually, in the maximum load power operation conditionillustrated in FIGS. 11A and 11B, the period TOFF is 2 microseconds, andthe period TON is 9 microseconds.

In FIGS. 11A to 11D, the resonance voltage Vcp is 700 Vp (FIG. 11C)during steady state operation, but rises up to 900 Vp (FIG. 11A) in amaximum load power operation condition. Meanwhile, the collector currentIcp is 4.5 Ap (FIG. 11D) in steady state operation, but rises up to 6.5Ap (FIG. 11B) in a maximum load power operation condition. The resonancevoltage Vcp and the collector current Icp in a maximum load poweroperation condition exhibit arise of approximately 20% to 30% from thelevels exhibited in steady state operation. Thus, switching element Q1must be able to withstand the voltage of 900 Vp. Actually, however, adevice having a voltage withstanding property against 1,200 V isselected for the switching element Q1 to provide an additionalappropriate margin against malfunction. A device having a voltagewithstanding property against 1,200 V is comparatively expensive andlarge in size.

In short, even if an over current limiting circuit, such as that shownin FIG. 10 is provided to suppress a possible application of too muchcurrent upon the application of a heavy load, a device having acorresponding high voltage withstanding property must be selected forthe switching element. Consequently, the circuit cannot be reduced incost or size.

Further, because the circuit construction shown in FIG. 10 does notinvolve particular control for a rise of the secondary side DC voltageupon start-up of the circuit, the rise time of the secondary side DCoutput voltage level is correspondingly short. Therefore, the powersupply circuit of FIG. 10 has a problem also in that a margin against amalfunction of the circuit upon startup cannot be provided.

DISCLOSURE OF THE INVENTION

It is therefore an object of the invention to provide an improvedswitching power supply circuit.

It is a further object of the invention to provide an improved switchingpower supply circuit wherein a switching element having a comparativelylow voltage withstanding property and a low current capacity can beselectively used and the switching power supply circuit can be producedwith low cost and with a small size.

Still other objects and advantages of the invention will in part beobvious and will in part be apparent from the specification and thedrawings.

Generally speaking, in accordance with the invention, an improvedswitching power supply circuit is provided, comprising a rectifiersmoother for receiving a commercial ac power supply as an input thereto,for producing a rectified smoothed voltage from the inputtedcoimimnercial ac power supply, and for outputting the rectified smoothedvoltage as a DC input voltage. A switcher including a switching elementfor receiving and switchably outputting the DC input voltage is alsoprovided. The switching element includes a self-excited oscillationdrive circuit for switchably driving the switching element in aself-excited manner, an insulating converter transformer fortransmitting the output of the switching element to a secondary windingside of the circuit, a primary winding side resonance circuit formedfrom at least a leakage inductance component including a primary windingof the insulating converter transformer and a capacitance of a resonancecapacitor for providing resonance type operation of the switchingelement. The switcher further includes a secondary winding sideresonance circuit formed from the leakage inductance component of thesecondary winding of the insulating converter transformer and thecapacitance of a secondary winding side resonance capacitor. A DC outputvoltage producer is also provided, including the secondary winding sideresonance circuit, for receiving an alternating voltage obtained at thesecondary winding of the insulating converter transformer as an inputthereto, and for performing a full-wave rectification operation for theinput alternating voltage to produce a secondary winding side DC outputvoltage. A switching frequency varying element is provided for varyingan inductance of the self-excited oscillation drive circuit in responseto a control current level supplied thereto to vary a switchingfrequency of the switching element. A constant voltage controller forvarying the control current in response to the level of the secondarywinding side DC output voltage and supplying the varied control currentto the switching frequency variation means to variably control theswitching frequency thereby to effect constant voltage control for thesecondary side DC output voltage in a feedback control type setup isalso provided. A switching frequency controller is provided forreceiving a start-up power supply upon or immediately after start-up ofthe switching power supply circuit as an input thereto, and forsupplying the control current of a predetermined level over apredetermined period of time after start-up in place of a controlcurrent supplied by the constant voltage controller to control theswitching frequency so that the switching frequency remains within apredetermined desired range.

For example, in a self-excited composite resonance switching converter,within a transient period until the secondary winding side DC outputvoltage is stabilized at its steady-state level after start-up of thepower supply, it is not possible to control the switching frequencyappropriately to stabilize the same using the standard switching controlused at steady-state. Therefore, the switching frequency controlleroperates upon start-up or immediately after start-up of the power supplycircuit so that the switching frequency thereof remains within apredetermined required range. In this manner, the switching frequency iscontrolled so that the secondary winding side DC output voltage levelmay be suppressed as in the switching power supply circuit of thepresent invention. Thus, a sudden rise of the secondary side DC outputvoltage upon start-up is suppressed, and a soft start-up is realized.

More specifically, in the switching power supply circuit according tothe invention, a composite resonance switching converter is providedwith a switching element that is driven by self-excited oscillation, andincludes a soft start-up circuit (switching frequency control means) sothat, upon start-up of a switching operation, the switching frequency israised up to a frequency substantially near to an upper limit of anallowable range of variation of the switching frequency to suppress asudden rise of the secondary winding side DC power output.

As a result, because the resonance voltage applied to the switchingelement, and therefore the switching output current (collector current)flowing to the switching element upon start-up of a switching operationare suppressed, a device having a comparatively low voltage withstandingproperty and having a comparatively low current capacity can beselectively used for the switching element. Consequently, the switchingpower supply circuit can be produced at a low cost and in a small size.

Further, it is not required for the switching power supply circuit ofthe invention to include a separate circuit for limiting over current tothe switching element. Consequently, power loss which might otherwiseoccur when using a separate current limiting circuit is eliminated.

Preferably, the switching power supply circuit in accordance with theinvention further comprises a tertiary winding wound at a position onthe primary winding of the insulating converter transformer spaced bymore than a predetermined physical distance from the secondary winding,and a rectifier smoothing circuit connected to the tertiary winding. Thestarting power supply which is supplied to the switching frequencycontroller is a DC voltage obtained by the rectifier smoothing circuit.According to this preferred start-up circuit, the soft start-up circuit(switching frequency controller) can be operated by the tertiary windingand the rectifier smoothing circuit which have a single independentconstruction.

Alternatively, the switching power supply circuit may further comprise astandby power supplier for receiving the commercial ac power supply asan input thereto and producing a DC voltage as a standby power supplyvoltage from the input commercial ac power supply. A switching startermay also be provided for making use of the standby power supply voltageto produce a switching starting signal which can be used to start theswitcher. The switching starting signal is used as the starting powersupply to be supplied to the switching frequency controller. With such aswitching power supply circuit, a tertiary winding as described aboveneed not be provided, and an insulating converter transformer having anordinary simple construction can be used for the insulating convertertransformer.

In the switching power supply circuit according to the presentinvention, the period within which the soft start-up circuit (switchingfrequency controller) operates depends upon a time constant provided bythe capacitance of, for example, a capacitor connected to the base of atransistor of the soft start-up circuit and a resistance component of acontrol winding to which control current is supplied. This signifiesthat the operation period of the soft start-up circuit can be setvariably if the capacitance of the capacitor or the resistance componentof the control winding is varied.

Thus, preferably the switching power supply circuit further comprises anoperation period setter for variably setting an operation period of theswitching frequency controller. Where the operation period of the softstart-up circuit is variably set in this manner, the time until thesecondary winding side output voltage rises to its steady level can beset arbitrarily. Consequently, the margin against a malfunction whenstarting the power supply can be increased advantageously.

The invention accordingly comprises the several steps and the relationof one or more of such steps with respect to each of the others, and theapparatus embodying features of construction, combination(s) of elementsand arrangement of parts that are adapted to effect such steps, all asexemplified in the following detailed disclosure, and the scope of theinvention will be indicated in the claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the invention, reference is made tothe following description and accompanying drawings, in which:

FIG. 1 is a circuit diagram depict a power supply circuit constructed inaccordance with the invention;

FIG. 2 is a side elevational sectional view showing a structure of aninsulating converter transformer of the invention provided in the powersupply circuit of FIG. 1;

FIGS. 3A and 3B are waveform diagrams illustrating a switching operationof the power supply circuit of FIG. 1;

FIG. 4 is a diagram comprising a soft start-up operation of the powersupply circuit of FIG. 1 of the invention and operation upon start-up ofa conventional power supply circuit;

FIG. 5 is a circuit diagram showing a construction of an alternativeembodiment of the invention;

FIG. 6 is a diagram view comparing a soft start-up operation of thepower supply circuit of FIG. 5 of the invention and operation uponstart-up of a standard power supply circuit;

FIG. 7 is a circuit diagram showing a construction of a standard powersupply circuit;

FIG. 8 is a sectional view showing a structure of a standard insulatingconverter transformer of the power supply circuit of FIG. 7;

FIGS. 9A and 9B are circuit diagrams illustrating operations of thestandard insulating converter transformer shown in FIG. 8 when themutual inductance is +M and −M, respectively;

FIG. 10 is a circuit diagram showing a construction of a standard powersupply circuit which includes an over current limitation circuit; and

FIGS. 11A to 11D are waveform diagrams illustrating switching operationof the standard power supply circuit shown in FIG. 10.

BEST MODE FOR CARRYING OUT THE INVENTION

Referring first to FIG. 1, a power supply circuit 100 constructed inaccordance with the invention is shown. Power supply circuit 100includes several components common with those of the standard powersupply apparatus described above with reference to FIG. 7. Descriptionof such common components is omitted herein to avoid redundancy, likeelements being designated by like reference numerals.

Power supply circuit 100 first includes a control circuit 1. Controlcircuit 1 includes a pair of resistors R3 and R4 connected in seriesbetween a DC output voltage E01 and a secondary winding side ground. Ashunt regulator Q3 is provided with a control terminal connected to ajunction (voltage dividing point) between resistors R3 and R4. The anodeof shunt regulator Q3 is grounded, and the cathode of shunt regulator Q3is connected to a line of a DC voltage E3 through a control winding NCof an orthogonal control transformer PRT, described below. The cathodeof shunt regulator Q3 is connected to the junction between resistors R3and R4 through a series connection of a capacitor C11 and a resistor R2.Further, a series connection circuit of a capacitor C3 and a resistor R5is connected in parallel to resistor R4.

Control circuit 1 functions as an error amplifier which receives DCoutput voltage E01 as a detection input thereto. In particular, avoltage obtained by voltage division of the DC output voltage E01 bymeans of resistors R3 and R4 is input to the control terminal of shuntregulator Q3. Accordingly, shunt regulator Q3 supplies current of alevel corresponding to the DC output voltage E01 to control winding NC.The level of the control current to flow through control winding NC istherefore variably controlled. Accordingly, the switching frequency ofswitching element Q1 is varied to execute constant voltage control asdescribed above with reference to FIG. 7.

Power supply circuit 100 further includes a soft start-up circuit 2 anda rectifier circuit for producing a DC voltage E3 for driving the softstart-up circuit 2. A construction of the rectifier circuit forproducing the DC voltage E3 will be first described.

The rectifier circuit includes a tertiary winding N3 wound on aninsulating converter transformer PIT, a diode D6, and a smoothingcapacitor C6 for half-wave rectifying an alternating voltage excited intertiary winding N3. Accordingly, DC voltage E3 is obtained acrosssmoothing capacitor C6. A resistor R7 is connected in parallel tosmoothing capacitor C6. DC voltage E3 across smoothing capacitor C6 issupplied as a detection voltage (operating power supply) to soft startupcircuit 2 through a Zener diode D7. Zener diode D7 is set, for example,in combination with resistor R7 connected in parallel to smoothingcapacitor C6, so as to become conducting with a reverse voltagecorresponding to 18 V.

Referring next o FIG. 2, a structure of an insulating convertertransformer PIT on which tertiary winding N3 is wound is shown. Portionsof the insulating converter transformer PIT shown in FIG. 2 areconstructed similarly to those in insulating converter transformer PITdescribed above in reference to FIG. 8. Thus, a gap G is formed betweenthe middle magnetic legs thereof so that a loose coupling having acoupling coefficient of, for example, approximately 0.85 may beobtained. Also, the mounting structures of primary winding N1 andsecondary winding N2 are similar to those in FIG. 8.

However, insulating converter transformer PIT shown in FIG. 2 isdifferent from the insulating converter transformer of FIG. 8 in thattertiary winding N3 is additionally wound, for example, as shown in FIG.2, at a position on the outer side (upper side) of a split bobbin Badjacent the primary winding N1 and farthest from the secondary windingN2. Alternatively, tertiary winding N3 may be wound at a position on thecenter side (lower side) of the split bobbin B farthest from thesecondary winding N2 as indicated as (N3) in FIG. 2. For tertiarywinding N3, for example, a triple insulating line is adopted.

Tertiary winding N3 is wound at a position farthest from secondarywinding N2 so that the coupling degree of tertiary winding N3 tosecondary winding N2 is minimized. If too much coupling is present, thelevel of DC voltage E3 upon start-up of the power supply may exceed asteady voltage as will be described below making reference to FIG. 4.Only if a coupling degree below a particular level can be obtained canthe tertiary winding N3 be wound at a particular position, for example,nearer to secondary winding N2 than the position shown in FIG. 2.

Soft start-up circuit 2 shown in FIG. 1 has the following construction.A pair of resistors R6 and R5′ are connected between the anode side ofZener diode D7 to which the DC voltage E3 is applied and the secondarywinding side ground. The base of a transistor Q5 (NPN) is connected tothe junction between resistors R6 and R5′.

The collector of transistor Q5 is connected to the base of a transistorQ4 through a resistor R7′. Further, a capacitor C4 for adjusting thebase potential applied to transistor Q4 is interposed in parallelbetween the base of transistor Q4 (PNP) and the secondary winding sideground. The emitter of transistor Q4 is connected to a junction betweencontrol winding NC of orthogonal control transformer PRT and the cathodeof shunt regulator Q3. The collector of transistor Q4 is connected tothe secondary winding side ground.

Operation of soft start-up circuit 2 will now be described makingreference to FIGS. 3A, 3B and 4. FIGS. 3A and 3B illustrate a resonancevoltage Vcp and a parallel collector current Icp corresponding, forexample, to a switching operation of switching element Q1 approximately20 ms after an AC switch SW is switched on in FIG. 1.

FIG. 4 illustrates level variations of the secondary side DC outputvoltage E0 (E01) and the DC voltage E3 upon start-up. In FIG. 4, thex-axis indicates the time after AC switch SW is switched on and they-axis indicates the voltage level. After AC switch SW is switched on,DC voltage E3 rises as shown in FIG. 4, and approximately after thelapse of 20 ms, a level (18 V or more) higher than 15 V, which is to bethe resulting steady state voltage level, is obtained. This operationarises from the tertiary winding N3 wound in such a manner as describedabove in FIG. 2.

When DC voltage E3 becomes 18 V or more as described above, the Zenerdiode D7 is rendered conducting. Consequently, DC voltage E3 is dividedby voltage dividing resistors R6 and R5′ in soft start-up circuit 2, andcurrent of a level corresponding to the divided voltage level flows tothe base of transistor Q5. Consequently, transistor Q5 is renderedconducting, and collector current of transistor Q5 flows as base currentto the base of transistor Q4 through the resistor R7′. Consequently,transistor Q4 is rendered conductive between the emitter and thecollector thereof At this time, secondary winding side DC output voltageE0 is in a transition period within which it rises toward 135 V which isthe steady state level thereof as shown in FIG. 4. Control circuit 1 isnot operating during this time.

Accordingly, when approximately 20 ms elapses after AC switch SW isswitched on, emitter current of transistor Q4 of soft start-up circuit 2flows to control winding NC of orthogonal control transformer PRT.Control circuit 1 of power supply circuit 100 of FIG. 1 controls so thatthe switching frequency rises as the level of the control currentflowing through control winding NC increases. Accordingly, as theemitter current of transistor Q4 of soft start-up circuit 2 flows tocontrol winding NC, control current of a sufficiently high level flowsto control winding NC. The switching frequency of switching element Q1is controlled so that approximately 20 ms after AC switch SW is switchedon, the switching frequency is adjusted, for example, up toapproximately 250 KHz which is the highest frequency within apredetermined allowable range of switching frequency variation.

Operation of switching element Q1 20 ms after AC switch SW is switchedon is illustrated in the waveform diagrams of FIGS. 3A and 3B. Inparticular, as can be seen from the waveforms of the resonance voltageVcp of FIG. 3A and the collector current Icp of FIG. 3B, switchingelement Q1 performs a switching operation in a cycle of the periodTOFF=period TON=2 microseconds. Because the switching operation isperformed with a high switching frequency (250 KHz) in this manner, theresonance voltage Vcp obtained when the switching element Q1 is off(within the period TOFF) is controlled to be approximately to 500 Vp asseen from FIG. 3A, and the parallel collector current Icp obtained whenswitching element Q1 is on (within the period TON) is 2 Ap.

Thereafter, charging of capacitor C4 is preformed in accordance with atime constant provided by the capacitance thereof Capacitor C4 isconnected to the base of transistor Q4 and a DC resistance component ofcontrol winding NC. This acts to raise the base potential of transistorQ4. As the base current level of transistor Q4 drops, the controlcurrent (emitter current) flowing through the control winding NCdecreases and the switching frequency of switching element Q1 graduallydrops.

In short, the power supply circuit of FIG. 1 performs control so thatapproximately 20 ms after AC switch SW is switched on by operation ofsoft start-up circuit 2, the switching frequency is raised compulsorily,and thereafter, the switching frequency is dropped gradually.Consequently, as can be recognized from comparison of the variations ofthe secondary side DC output voltage E0 by the power supply circuit ofFIG. 1 according to the present invention indicated by a solid line inFIG. 4 and the power supply circuit of FIG. 10 indicated by a brokenline in FIG. 4, the power supply circuit of FIG. 1 operates so that thetime until the secondary side DC output voltage E0 rises to itssteady-state level (135 V) upon start-up is longer.

From FIG. 4, it can be seen that power supply circuit 100 of FIG. 1 isset so that the rise time of the secondary winding side DC outputvoltage E0 from power supply circuit 100 of FIG. 1 is approximately fivetimes the rise time of the power supply circuit of FIG. 10. Inparticular, while the power supply circuit of FIG. 10 reaches itssteady-state level at approximately 20 ms after the AC switch SW isswitched on, power supply circuit 100 of FIG. 1 is controlled so thatits steady-state level is reached approximately 100 ms after AC switchSW is switched on. Because control circuit 1 operates properly whensecondary winding side DC output voltage E0 returns to its steady levelapproximately 100 ms after AC switch SW is switched on, the switchingfrequency control operation for the switching element Q1 is passed fromsoft start-up circuit 2 to control circuit 1. Further, at this point oftime, because DC voltage E3 returns to 15 V, its steady level, Zenerdiode D7 is rendered non-conducting. Accordingly, operation of softstart-up circuit 2 is also stopped.

In the power supply circuit according to the invention, a soft start-upoperation upon start-up of a power supply is realized. According to thissoft start-up operation, the resonance voltage V_(cp) upon start-up iscontrolled to 500 Vp with the power supply circuit of FIG. 1, while in aconventional power supply circuit, the resonance voltage V_(cp) uponstart-up is 900 VP to 1,000 V_(p) with the power supply circuit of FIG.10. Also collector current level upon start-up is controlled to be 2 Apaccording to the power supply circuit of FIG. 1 of the invention whilethis level may rise to 6.5 Ap with the power supply circuit of FIG. 10.

Consequently, a device having a voltage withstanding property of only700 V and having a comparatively small current capacity can be employedfor switching element Q1 of the power supply circuit of FIG. 1. Aninexpensive device of a comparatively small size can therefore beproduced. Additionally, the capacitor Cr on the primary winding side ofthe circuit can employ a device able to withstand only approximately 800V.

In the power supply circuit of FIG. 1, the time for transition after asoft startup operation until reaching steady state operation dependsupon the time constant provided by capacitor C4 in soft start-up circuit2, and control winding NC. The time for transition after a soft start-upoperation until reaching a steady-state operation can be set arbitrarilyby adjusting at least one of, for example, the capacitance of capacitorC4 and the DC resistance component (turn number) of control winding NC.Accordingly, a malfunction margin upon start-up can be increased byadjustment of the time constant. Because the DC resistance component(turn number) of control winding NC is principally set depending upon acontrol characteristic of the orthogonal control transformer PRT,preferably the capacitance of capacitor C4 is varied.

FIG. 5 shows another power supply circuit 500 constructed in accordancewith an alternative embodiment of the invention. Power supply circuit500 is a modification of the power supply circuit 100 described inreference to FIG. 1 in that it additionally includes a standby powersupply circuit 4. Power supply circuit 500 shown in FIG. 5 is thereforepractically more advantageous than power supply circuit 100 of FIG. 1.

Standby power supply circuit 4 of FIG. 5 includes a standby transformerSBT which also receives ac input voltage VAC. Standby transformer SBTincludes a secondary winding, a pair of rectifier diodes D11 and D12 anda smoothing capacitor C03 connected in such a manner as shown in FIG. 5to form a center tap type full-wave rectifier circuit. Standbytransformer SBT produces a rectified smoothed voltage of, for example, 7V using ac input voltage VAC input to the primary side thereof Therectified smoothed voltage of 7 V is supplied to a regulator RG. Theregulator RG stabilizes the input voltage of 7 V and outputs astabilized voltage of 5 V. The stabilized DC voltage of 5 V is obtainedas a voltage across a smoothing capacitor C04. The DC voltage of 5 V issupplied as a standby power supply voltage and as an operation powersupply, for example, for a microcomputer, an IC for reception ofinfrared rays and so forth included in the apparatus. DC voltage of 5 Vis supplied also as an operation power supply for a relay drive circuit3. Relay drive circuit 3 is provided to control an on/off operation ofAC switch SW with an on/off control signal for the main power supplycircuit output under the control of, for example, the microcomputer. Tothis end, an electromagnetic relay RY of the relay drive circuit 3operates in association with an on/off operation of AC switch SW.

In power supply circuit 500, the on/off control signal for the mainpower supply circuit is a DC voltage having a level higher than apredetermined level to drive relay drive circuit 3 and soft start-upcircuit 2 in such a manner as described below. Then, when the main powersupply circuit is to be turned on, the DC voltage mentioned above isoutput, but when the main power supply circuit is to be turned off, theoutput of the DC voltage is stopped. The DC voltage as the on/offcontrol signal for the main power supply circuit is obtained, though notshown in FIG. 5, by the microcomputer making use of the DC voltage of 5V produced by standby power supply circuit 4.

In relay drive circuit 3, a pair of voltage dividing resistors R21 andR22 are interposed between the line for the on/off control signal forthe main power supply circuit and the secondary winding side ground. Thebase of a transistor Q2 is connected to a junction between voltagedividing resistors R21 and R22. A capacitor C10 is connected between thebase of the transistor Q2 and the secondary winding side ground.

The collector of transistor Q2 is connected to the DC voltage line of 5V obtained from standby power supply circuit 4 through electromagneticrelay RY. The emitter of transistor Q2 is grounded to the secondarywinding side ground. A protecting diode D8 is connected in a directionshown in FIG. 5 to electromagnetic relay RY.

Further, in power supply circuit 500, the on/off control signal for themain power supply circuit is input as a detection voltage to softstart-up circuit 2. In soft start-up circuit 2 shown in FIG. 5, the DCvoltage level as the on/off control signal is divided by the voltagedividing resistors R6 and R5′ to perform detection thereby to effectconduction control of transistors Q5 and Q4. Accordingly, in powersupply circuit 500, the tertiary winding N3 used in the power supplycircuit of FIG. 1 is not required. Consequently, the insulatingconverter transformer PIT of FIG. 8 can be used as the insulatingconverter transformer in power supply circuit 500.

A diode D5 shown in FIG. 5 forms a path from the power supply foroperating control circuit 1 upon soft start-up. The anode of diode D5 isconnected to the DC voltage line of 5 V of the standby power supplycircuit 4. The cathode of diode D5 is connected to the cathode of shuntregulator Q3 through control winding NC. A diode D4 forms a path fromthe power supply for operating control circuit 1 during steady operationafter soft start-up. The anode of diode D4 is connected to the line forthe secondary winding side DC output voltage E02 of the main powersupply circuit while the cathode of diode D4 is connected to the cathodeof shunt regulator Q3 through control winding NC. During use, if a DCvoltage is output as the on/off control signal for turning the mainpower supply on, then transistor Q2 of relay drive circuit 3 is renderedconductive, thereby to drive electromagnetic relay RY. Consequently, theAC switch SW is switched on, and power supply to the main power supplyside is started.

The DC voltage as the on/off control signal is input also to the seriesconnection circuit of resistors R6 and R5′ of soft start-up circuit 2.Consequently, transistors Q5 and Q4 are rendered conducting, and controlcurrent flows to the control winding NC as described above.

The switching frequency of switching element Q1 is controlled so that itrises substantially up to the highest frequency level (250 KHz) withinan allowable variation control range in a similar manner as in the powersupply circuit of FIG. 1 described above. Thus, secondary side DC outputvoltages E0 (DC output voltage E01 and E02) do not reach their steadylevels too quickly with a steep slope of increase. The controllingcondition in this instance is shown in FIG. 6. Level variations of thesecondary winding side DC output voltages E0 (DC output voltage E01 andE02) upon start-up are shown. As is seen from FIG. 6, a time ofapproximately 100 ms is required after AC switch SW is switched on untilthe DC output voltages E01 and E02 reach their respective steady-statelevels (here, DC output voltage E01=135 V and E02=15 V).

An operation similar to that described above with reference to thewaveform diagrams of FIGS. 3A and 3B is obtained as the switchingoperation of switching element Q1 immediately after startup.Accordingly, a device able to withstanding a voltage of 700 V and havinga comparatively small current capacity can be employed for switchingelement Q1. Also a device able to withstand a voltage of approximately800 V can be selectively used for parallel resonance capacitor Cr.Further, by adjusting at least one of the capacitance of capacitor C4and the DC resistance component (turn number) of control winding NC, amalfunction margin upon starting of power supply can be set similarly asin the power supply circuit of FIG. 1.

The switching power supply circuit of the present invention is notlimited to the specific forms described hereinabove but may have variousforms. For example, detailed constructions of the control circuit 1,soft start-up circuit 2, relay drive circuit 3 and so forth of the powersupply apparatus described hereinabove with reference to FIGS. 1 and 5may be modified suitably.

Further, while also a push-pull type voltage resonance converter whereintwo switching elements are turned on/off alternately is known, thepresent invention can be applied also to a self-excited voltageresonance converter which employs the push-pull system.

Furthermore, while also a composite resonance switching converterwherein a resonance capacitor is connected in series to the secondarywinding of insulating converter transformer PIT to form a secondary sideseries resonance circuit has been proposed by the assignee of thepresent application, the present invention can be applied also to acomposite switching converter of the type just described.

Further, while a voltage resonance converter is used in the power supplycircuit described hereinabove with reference to FIGS. 1 and 5, thepresent invention can be applied also to a power supply circuit in whicha current resonance converter wherein stabilization is performed byself-excited switching frequency control is provided on the primaryside.

It will thus be seen that the objects set forth above, among those madeapparent from the preceding description, are efficiently attained and,because certain changes may be made in carrying out the above method andin the construction(s) set forth without departing from the spirit andscope of the invention, it is intended that all matter contained in theabove description and shown in the accompanying drawings shall beinterpreted as illustrative and not in a limiting sense.

It is also to be understood that the following claims are intended tocover all of the generic and specific features of the invention hereindescribed and all statements of the scope of the invention which, as amatter of language, might be said to fall therebetween.

INDUSTRIAL APPLICABILITY

As described above, a switching power supply circuit according to thepresent invention allows selective use of a switching element having acomparatively low voltage withstanding property and a low currentcapacity and can be produced in a low lost and with a small size. Uponstarting of power supply, the switching frequency of the switchingelement is raised by a soft starting circuit to increase a period oftime until a secondary side DC output voltage reaches a steady levelthereof As a result, the levels of a resonance voltage and a collectorcurrent obtained when the switching element starts its switchingoperation are suppressed.

What is claimed is:
 1. A switching power supply circuit, comprising:rectifier smoothing means for receiving a commercial ac power supply asan input and producing and outputting a rectified smoothed DC voltage;switching means for providing a switched output, said switching meanscomprising: a switching element for receiving and switchably outputtingthe DC voltage; a self-excited oscillation drive circuit for switchablydriving said switching element in a self-excited manner; an insulatingconverter transformer for transmitting the output of said switchingmeans to a secondary winding side of said switching power supplycircuit; a primary winding side resonance circuit formed at least from aleakage inductance component including a primary winding of saidinsulating converter transformer and a capacitance of a resonancecapacitor, said primary winding side resonance circuit operating saidswitching means in a resonance type operation; and a secondary windingside resonance circuit formed on the secondary winding side of saidswitching power supply circuit, including a leakage inductance componentof a secondary winding of said insulating converter transformer and acapacitance of a secondary side resonance capacitor; DC output voltageproduction means, including said secondary side resonance circuit, forreceiving an alternating voltage obtained at said secondary winding ofsaid insulating converter transformer as an input thereto and performinga full-wave rectification operation for the input alternating voltage toproduce a secondary winding side DC output voltage; switching frequencyvariation means for varying an inductance of said self excitedoscillation drive circuit in response to a control current levelsupplied thereto to vary a switching frequency of said switchingelement; constant voltage control means for varying the control currentin response to the level of the secondary winding side DC output voltageand supplying the varied control current to said switching frequencyvariation means to variably control the switching frequency thereby toeffect constant voltage control for the secondary side DC outputvoltage; and switching frequency control means for receiving a start-uppower supply obtained upon, or immediately after start up of saidswitching power supply circuit as an input thereto and for supplying acontrol current of a predetermined level over a predetermined period oftime after start-up in place of said constant voltage control means tocontrol the switching frequency so that the switching frequency remainswithin a predetermined range.
 2. The switching power supply circuitaccording to claim 1, further comprising: a tertiary winding wound at aposition on said primary winding of said insulating convertertransformer spaced by more than a predetermined physical distance fromsaid secondary winding; and a rectifier smoothing circuit connected tosaid tertialy winding, the start-up power supply which is supplied tosaid switching frequency control means being a DC voltage obtained bysaid rectifier smoothing circuit.
 3. The switching power supply circuitas claimed in claim 1, further comprising: standby power supply meansfor receiving the commercial ac power supply as an input thereto andproducing a DC voltage as a standby power supply voltage from the inputcommercial ac power supply; and switching start-up means for making useof the standby power supply voltage to produce a switching start-upsignal which can be used to start said switching means, the switchingstart-up signal being used as the start-up power supply to be suppliedto said switching frequency control means.
 4. The switching power supplycircuit according to claim 1, further comprising operation periodsetting means for variably setting an operation period of said switchingfrequency control means.
 5. A switching power supply circuit,comprising: a rectifier smoother for receiving a commercial ac powersupply as an input and producing and outputting a rectified smoothed DCvoltage; a switch for providing a switched output, said switchcomprising: a switching element for receiving and switchably outputtingthe DC voltage; and a self-excited oscillation drive circuit forswitchably driving said switching element in a self-excited manner; aninsulating converter transformer for transmitting the output of saidswitching means to a secondary winding side of said switching powersupply circuit; a primary winding side resonance circuit formed at leastfrom a leakage inductance component including a primary winding of saidinsulating converter transformer and a capacitance of a resonancecapacitor, said primary winding side resonance circuit operating saidswitch in a resonance type operation; and a secondary winding sideresonance circuit formed on the secondary winding side of said switchingpower supply circuit, including a leakage inductance component of asecondary winding of said insulating converter transformer and acapacitance of a secondary side resonance capacitor; DC output voltageproducer, including said secondary side resonance circuit, for receivingan alternating voltage obtained at said secondary winding of saidinsulating converter transformer as an input thereto and performing afull-wave rectification operation for the input alternating voltage toproduce a secondary winding side DC output voltage; a switchingfrequency varier for varying an inductance of said self-excitedoscillation drive circuit in response to a control current levelsupplied thereto to vary a switching frequency of said switchingelement; a constant voltage controller for varying the control currentin response to the level of the secondary winding side DC output voltageand supplying the varied control current to said switching frequencyvarier means to variably control the switching frequency thereby toeffect constant voltage control for the secondary side DC outputvoltage; and a switching frequency controller for receiving a start-uppower supply obtained upon, or immediately after start up of saidswitching power supply circuit as an input thereto and for supplying acontrol current of a predetermined level over a predetermined period oftime after start-up in place of said constant voltage controller tocontrol the switching frequency so that the switching frequency remainswithin a predetermined range.
 6. The switching power supply circuitaccording to claim 5, further comprising: a tertiary winding wound at aposition on said primary winding of said insulating convertertransformer spaced by more than a predetermined physical distance fromsaid secondary winding; and a rectifier smoothing circuit connected tosaid tertiary winding, the start-up power supply which is supplied tosaid switching frequency control means being a DC voltage obtained bysaid rectifier smoothing circuit.
 7. The switching power supply circuitas claimed in claim 5, farther comprising: a standby power supplier forreceiving the commercial ac power supply as an input thereto andproducing a DC voltage as a standby power supply voltage from the inputcommercial ac power supply; and a start-up switcher for making use ofthe standby power supply voltage to produce a switching start-up signalwhich can be used to start said switcher, the switching start-up signalbeing used as the start-up power supply to be supplied to said switchingfrequency controller.
 8. The switching power supply circuit according toclaim 5, further comprising an operation period setter means forvariably setting an operation period of said switching frequencycontroller.
 9. A switching power supply method, comprising the steps of:receiving a commercial ac power supply by a rectifier smoother as aninput; producing and outputting a rectified smoothed DC voltage;providing a switched output by a switcher, said switcher comprising: aswitching element for receiving and switchably outputting the DCvoltage; a self-excited oscillation drive circuit for switchably drivingsaid switching element in a self-excited manner; an insulating convertertransformer for transmitting the output of said switching means to asecondary winding side of said switching power supply circuit; a primarywinding side resonance circuit formed at least from a leakage inductancecomponent including a primary winding of said insulating convertertransformer and a capacitance of a resonance capacitor, said primarywinding side resonance circuit operating said switching means in aresonance type operation; and a secondary winding side resonance circuitformed on the secondary winding side of said switching power supplycircuit, including a leakage inductance component of a secondary windingof said insulating converter transformer and a capacitance of asecondary side resonance capacitor; receiving an alternating voltage asan input to a DC output voltage producer, including said secondary sideresonance circuit, obtained at said secondary winding of said insulatingconverter transformer; performing a full-wave rectification operationfor the input alternating voltage to produce a secondary winding side DCoutput voltage; varying an inductance of said self-excited oscillationdrive circuit by a switching frequency varier in response to a controlcurrent level supplied thereto to vary a switching frequency of saidswitching element; varying the control current in response to the levelof the secondary winding side DC output voltage; supplying the variedcontrol current to said switching frequency varier to variably controlthe switching frequency thereby to effect constant voltage control forthe secondary side DC output voltage; and receiving a start-up powersupply by a switching frequency controller obtained upon or immediatelyafter start up of said switching power supply circuit as an inputthereto and for supplying a control current of a predetermined levelover a predetermined period of time after start-up in place of saidsupplied control current to control the switching frequency so that itremains within a predetermined range.
 10. The method according to claim9, further comprising the steps of: providing a tertiary winding at aposition on said primary winding of said insulating convertertransformer spaced by more than a predetermined physical distance fromsaid secondary winding; and connecting a rectifier smoothing circuit tosaid tertiary winding, whereby the start-up power supply supplied tosaid switching frequency control means is a DC voltage obtained by saidrectifier smoothing circuit.
 11. The method of claim 9, furthercomprising the steps of receiving the commercial ac power supply by astandby power supplier as an input thereto; producing a DC voltage as astandby power supply voltage from the input commercial ac power supply;and producing a switching start-up signal according to the standby powersupply voltage which can be used to start said switcher, the switchingstart-up signal being used as the start-up power supply to be suppliedto said switching frequency controller.
 12. The method of claim 9,further comprising the step of variably setting an operation period ofsaid switching frequency controller.
 13. A switching power supplycircuit including a resonance type converter, comprising: switchingfrequency control means for receiving a start-up power supply obtainedupon, or immediately after start-up of said switching power supplycircuit as an input obtained thereto; and means for supplying a controlcurrent of a predetermined level over a predetermined period of timeafter start-up in place of a voltage control signal to control aswitching frequency of a switching element so that the switchingfrequency remains within a predetermined range, whereby said switchingfrequency is raised immediately after start-up and gradually decreasedthereafter.
 14. A switching power supply circuit including a resonancetype converter, comprising: a switching frequency controller forreceiving a start-up power supply obtained upon, or immediately afterstart-up of said switching power supply circuit as an input obtainedthereto; and a supplier for supplying a control current of apredetermined level over a predetermined period of time after start-upin place of a voltage control signal to control a switching frequency ofa switching element so that the switching frequency remains within apredetermined range, whereby said switching frequency is raisedimmediately after start-up and gradually decreased thereafter.
 15. Aswitching power supply method employing a resonance type converter,comprising the steps of: receiving as an input a start-up power supplyobtained on, or immediately after start-up; and supplying a controlcurrent of a predetermined level over a predetermined period of timeafter start-up in place of a voltage control signal to control aswitching frequency of a switching element so that the switchingfrequency remains within a predetermined range, whereby said switchingfrequency is raised immediately after start-up and gradually decreasedthereafter.